步進電機控制系統(tǒng)外文翻譯

1、步進電機的振蕩、不穩(wěn)定以及控制摘要：本文介紹了一種分析永磁步進電機不穩(wěn)定性的新穎方法。結果表明，該種電機有兩種類型的不穩(wěn)定現(xiàn)象：中頻振蕩和高頻不穩(wěn)定性。非線性分叉理論是用來說明局部不穩(wěn)定和中頻振蕩運動之間的關系。一種新型的分析介紹了被確定為高頻不穩(wěn)定性的同步損耗現(xiàn)象。在相間分界線和吸引子的概念被用于導出數(shù)量來評估高頻不穩(wěn)定性。通過使用這個數(shù)量就可以很容易地估計高頻供應的穩(wěn)定性。此外，還介紹了穩(wěn)定性理論。廣義的方法給出了基于反饋理論的穩(wěn)定問題的分析。結果表明，中頻穩(wěn)定度和高頻穩(wěn)定度可以提高狀態(tài)反饋。關鍵詞：步進電機，不穩(wěn)定，非線性，狀態(tài)反饋。1. 介紹步進電機是將數(shù)字脈沖輸入轉換為模擬角度輸出的

2、電磁增量運動裝置。其內(nèi)在的步進能力允許沒有反饋的精確位置控制。 也就是說，他們可以在開環(huán)模式下跟蹤任何步階位置，因此執(zhí)行位置控制是不需要任何反饋的。步進電機提供比直流電機每單位更高的峰值扭矩;此外，它們是無電刷電機，因此需要較少的維護。所有這些特性使得步進電機在許多位置和速度控制系統(tǒng)的選擇中非常具有吸引力，例如如在計算機硬盤驅(qū)動器和打印機，代理表，機器人中的應用等.盡管步進電機有許多突出的特性，他們?nèi)栽馐苷袷幓虿环€(wěn)定現(xiàn)象。這種現(xiàn)象嚴重地限制其開環(huán)的動態(tài)性能和需要高速運作的適用領域。 這種振蕩通常在步進率低于1000脈沖/秒的時候發(fā)生，并已被確認為中頻不穩(wěn)定或局部不穩(wěn)定1，或者動態(tài)不穩(wěn)定2。此外

3、，步進電機還有另一種不穩(wěn)定現(xiàn)象，也就是在步進率較高時，即使負荷扭矩小于其牽出扭矩，電動機也常常不同步。該文中將這種現(xiàn)象確定為高頻不穩(wěn)定性，因為它以比在中頻振蕩現(xiàn)象中發(fā)生的頻率更高的頻率出現(xiàn)。高頻不穩(wěn)定性不像中頻不穩(wěn)定性那樣被廣泛接受，而且還沒有一個方法來評估它。中頻振蕩已經(jīng)被廣泛地認識了很長一段時間，但是，一個完整的了解還沒有牢固確立。這可以歸因于支配振蕩現(xiàn)象的非線性是相當困難處理的。大多數(shù)研究人員在線性模型基礎上分析它1。盡管在許多情況下，這種處理方法是有效的或有益的，但為了更好地描述這一復雜的現(xiàn)象，在非線性理論基礎上的處理方法也是需要的。例如，基于線性模型只能看到電動機在某些供應頻率下轉向

4、局部不穩(wěn)定，并不能使被觀測的振蕩現(xiàn)象更多深入。事實上，除非有人利用非線性理論，否則振蕩不能評估。因此，在非線性動力學上利用被發(fā)展的數(shù)學理論處理振蕩或不穩(wěn)定是很重要的。值得指出的是，Taft和Gauthier3，還有Taft和Harned4使用的諸如在振蕩和不穩(wěn)定現(xiàn)象的分析中的極限環(huán)和分界線之類的數(shù)學概念，并取得了關于所謂非同步現(xiàn)象的一些非常有啟發(fā)性的見解。盡管如此，在這項研究中仍然缺乏一個全面的數(shù)學分析。本文一種新的數(shù)學分被開發(fā)了用于分析步進電機的振動和不穩(wěn)定性。本文的第一部分討論了步進電機的穩(wěn)定性分析。結果表明，中頻振蕩可定性為一種非線性系統(tǒng)的分叉現(xiàn)象（霍普夫分叉）。本文的貢獻之一是將中頻振

5、蕩與霍普夫分叉聯(lián)系起來，從而霍普夫理論從理論上證明了振蕩的存在性。高頻不穩(wěn)定性也被詳細討論了，并介紹了一種新型的量來評估高頻穩(wěn)定。這個量是很容易計算的，而且可以作為一種標準來預測高頻不穩(wěn)定性的發(fā)生。在一個真實電動機上的實驗結果顯示了該分析工具的有效性。本文的第二部分通過反饋討論了步進電機的穩(wěn)定性控制。一些設計者已表明，通過調(diào)節(jié)供應頻率 5 ，中頻不穩(wěn)定性可以得到改善。特別是Pickup和Russell 6，7都在頻率調(diào)制的方法上提出了詳細的分析。在他們的分析中，雅可比級數(shù)用于解決常微分方程和一組數(shù)值有待解決的非線性代數(shù)方程組。此外，他們的分析負責的是雙相電動機，因此，他們的結論不能直接適用于我

6、們需要考慮三相電動機的情況。在這里，我們提供一個沒有必要處理任何復雜數(shù)學的更簡潔的穩(wěn)定步進電機的分析。在這種分析中，使用的是d-q模型的步進電機。由于雙相電動機和三相電動機具有相同的d-q模型，因此，這種分析對雙相電動機和三相電動機都有效。迄今為止，人們僅僅認識到用調(diào)制方法來抑制中頻振蕩。本文結果表明，該方法不僅對改善中頻穩(wěn)定性有效，而且對改善高頻穩(wěn)定性也有效。2. 動態(tài)模型的步進電機本文件中所考慮的步進電機由一個雙相或三相繞組的跳動定子和永磁轉子組成。一個極對三相電動機的簡化原理如圖1所示。步進電機通常是由被脈沖序列控制產(chǎn)生矩形波電壓的電壓源型逆變器供給的。這種電動機用本質(zhì)上和同步電動機相同

7、的原則進行作業(yè)。步進電機主要作業(yè)方式之一是保持提供電壓的恒定以及脈沖頻率在非常廣泛的范圍上變化。在這樣的操作條件下，振動和不穩(wěn)定的問題通常會出現(xiàn)。圖1.三相電動機的圖解模型 用qd框架參考轉換建立了一個三相步進電機的數(shù)學模型 。下面給出了三相繞組電壓方程va = Ria + L*dia /dt M*dib/dt M*dic/dt + dpma/dt ,vb = Rib + L*dib/dt M*dia/dt M*dic/dt + dpmb/dt ,vc = Ric + L*dic/dt M*dia/dt M*dib/dt + dpmc/dt , (1) 其中R和L分別是相繞組的電阻和感應線圈，

8、并且M是相繞組之間的互感線圈。pma, pmb and pmc 是應歸于永磁體 的相的磁通，且可以假定為轉子位置的正弦函數(shù)如下pma = 1 sin(N),pmb = 1 sin(N 2 QUOTE /3),pmc = 1 sin(N - 2 QUOTE /3), (2)其中N是轉子齒數(shù)。本文中強調(diào)的非線性由上述方程所代表，即磁通是轉子位置的非線性函數(shù)。使用Q ,d轉換，將參考框架由固定相軸變換成隨轉子移動的軸（參見圖2）。矩陣從a,b,c框架轉換成q,d框架變換被給出了8 (3)例如，給出了q,d參考里的電壓 (4)在a,b,c參考中，只有兩個變量是獨立的（ia + ib + ic = 0）

9、，因此，上面提到的由三個變量轉化為兩個變量是允許的。在電壓方程（1）中應用上述轉換，在q,d框架中獲得轉換后的電壓方程為vq = Riq + L1*diq/dt + NL1id + N1,vd = Rid + L1*did/dt NL1iq, (5) 圖2，a,b,c和d,q參考框架其中L1 = L + M，且是電動機的速度。有證據(jù)表明,電動機的扭矩有以下公式T = 3/2N1iq . (6)轉子電動機的方程為J*d/dt = 3/2*N1iq Bf Tl , (7) 如果Bf是粘性摩擦系數(shù)，和Tl代表負荷扭矩（在本文中假定為恒定）。為了構成完整的電動機的狀態(tài)方程，我們需要另一種代表轉子位置的

10、狀態(tài)變量。為此，通常使用滿足下列方程的所謂的負荷角8D/dt = 0 , (8) 其中0是電動機的穩(wěn)態(tài)轉速。方程（5），（7），和（8）構成電動機的狀態(tài)空間模型，其輸入變量是電壓vq和vd.如前所述，步進電機由逆變器供給，其輸出電壓不是正弦電波而是方波。然而，由于相比正弦情況下非正弦電壓不能很大程度地改變振蕩特性和不穩(wěn)定性（如將在第3部分顯示的，振蕩是由于電動機的非線性），為了本文的目的我們可以假設供給電壓是正弦波。根據(jù)這一假設，我們可以得到如下的vq和vdvq = Vmcos(N) ,vd = Vmsin(N) , (9) 其中Vm是正弦波的最大值。上述方程，我們已經(jīng)將輸入電壓由時間函數(shù)轉變

11、為狀態(tài)函數(shù)，并且以這種方式我們可以用自控系統(tǒng)描繪出電動機的動態(tài)，如下所示。這將有助于簡化數(shù)學分析。根據(jù)方程（5），（7），和（8），電動機的狀態(tài)空間模型可以如下寫成矩陣式 = F(X,u) = AX + Fn(X) + Bu , (10) 其中X = iq id T, u = 1 Tl T 定義為輸入，且1 = N0 是供應頻率。輸入矩陣B被定義為矩陣A是F(.)的線性部分，如下Fn(X)代表了F(.)的線性部分，如下輸入端u獨立于時間，因此，方程（10）是獨立的。在F(X,u)中有三個參數(shù)，它們是供應頻率1，電源電壓幅度Vm和負荷扭矩Tl。這些參數(shù)影響步進電機的運行情況。在實踐中，通常用這樣

12、一種方式來驅(qū)動步進電機，即用因指令脈沖而變化的供應頻率1來控制電動機的速度，而電源電壓保持不變。因此，我們應研究參數(shù)1的影響。3.分叉和中頻振蕩，設=0，得出方程（10）的平衡且是它的相角， = arctan(1L1/R) . (16) 方程（12）和（13）顯示存在著多重均衡，這意味著這些平衡永遠不能全局穩(wěn)定。人們可以看到，如方程（12）和（13）所示有兩組平衡。第一組由方程（12）對應電動機的實際運行情況來代表。第二組由方程（13）總是不穩(wěn)定且不涉及到實際運作情況來代表。在下面，我們將集中精力在由方程（12）代表的平衡上。 附件2：外文原文 Oscillation, Instability

13、 and Control of Stepper MotorsLIYU CAO and HOWARD M. SCHWARTZDepartment of Systems and Computer Engineering, Carleton University, 1125 Colonel By Drive,Ottawa, ON K1S 5B6, Canada(Received: 18 February 1998; accepted: 1 December 1998)Abstract. A novel approach to analyzing instability in permanent-ma

14、gnet stepper motors is presented. It is shown that there are two kinds of unstable phenomena in this kind ofmotor: mid-frequency oscillation and high-frequency instability. Nonlinear bifurcation theory is used to illustrate the relationship between local instability and midfrequencyoscillatory motio

15、n. A novel analysis is presented to analyze the loss of synchronism phenomenon, which is identified as high-frequency instability. The concepts of separatrices and attractors in phase-space are used to derive a quantity to evaluate the high-frequency instability. By using this quantity one can easil

16、y estimate the stability for high supply frequencies. Furthermore, a stabilization method is presented. A generalized approach to analyze the stabilization problem based on feedback theory is given. It is shown that the mid-frequency stabilityand the high-frequency stability can be improved by state

17、 feedback. Keywords: Stepper motors, instability, nonlinearity, state feedback.1. IntroductionStepper motors are electromagnetic incremental-motion devices which convert digital pulse inputs to analog angle outputs. Their inherent stepping ability allows for accurate position control without feedbac

18、k. That is, they can track any step position in open-loop mode, consequently no feedback is needed to implement position control. Stepper motors deliver higher peak torque per unit weight than DC motors; in addition, they are brushless machines and therefore require less maintenance. All of these pr

19、operties have made stepper motors a very attractive selection in many position and speed control systems, such as in computer hard disk drivers and printers, XY-tables, robot manipulators, etc.Although stepper motors have many salient properties, they suffer from an oscillation or unstable phenomeno

20、n. This phenomenon severely restricts their open-loop dynamic performance and applicable area where high speed operation is needed. The oscillation usually occurs at stepping rates lower than 1000 pulse/s, and has been recognized as a mid-frequency instability or local instability 1, or a dynamic in

21、stability 2. In addition, there is another kind of unstable phenomenon in stepper motors, that is, the motors usually lose synchronism at higher stepping rates, even though load torque is less than their pull-out torque. This phenomenon is identified as high-frequency instability in this paper, beca

22、use it appears at much higher frequencies than the frequencies at which the mid-frequency oscillation occurs. The high-frequency instability has not been recognized as widely as mid-frequency instability, and there is not yet a method to evaluate it.Mid-frequency oscillation has been recognized wide

23、ly for a very long time, however, a complete understanding of it has not been well established. This can be attributed to the nonlinearity that dominates the oscillation phenomenon and is quite difficult to deal with.384 L. Cao and H. M. SchwartzMost researchers have analyzed it based on a linearize

24、d model 1. Although in many cases, this kind of treatments is valid or useful, a treatment based on nonlinear theory is needed in order to give a better description on this complex phenomenon. For example, based on a linearized model one can only see that the motors turn to be locally unstable at so

25、me supplyfrequencies, which does not give much insight into the observed oscillatory phenomenon. In fact, the oscillation cannot be assessed unless one uses nonlinear theory.Therefore, it is significant to use developed mathematical theory on nonlinear dynamics to handle the oscillation or instabili

26、ty. It is worth noting that Taft and Gauthier 3, and Taft and Harned 4 used mathematical concepts such as limit cycles and separatrices in the analysis of oscillatory and unstable phenomena, and obtained some very instructive insights into the socalled loss of synchronous phenomenon. Nevertheless, t

27、here is still a lack of a comprehensive mathematical analysis in this kind of studies. In this paper a novel mathematical analysis is developed to analyze the oscillations and instability in stepper motors.The first part of this paper discusses the stability analysis of stepper motors. It is shown t

28、hat the mid-frequency oscillation can be characterized as a bifurcation phenomenon (Hopf bifurcation) of nonlinear systems. One of contributions of this paper is to relate the midfrequency oscillation to Hopf bifurcation, thereby, the existence of the oscillation is provedtheoretically by Hopf theor

29、y. High-frequency instability is also discussed in detail, and a novel quantity is introduced to evaluate high-frequency stability. This quantity is very easyto calculate, and can be used as a criteria to predict the onset of the high-frequency instability. Experimental results on a real motor show

30、the efficiency of this analytical tool.The second part of this paper discusses stabilizing control of stepper motors through feedback. Several authors have shown that by modulating the supply frequency 5, the midfrequencyinstability can be improved. In particular, Pickup and Russell 6, 7 have presen

31、ted a detailed analysis on the frequency modulation method. In their analysis, Jacobi series was used to solve a ordinary differential equation, and a set of nonlinear algebraic equations had to be solved numerically. In addition, their analysis is undertaken for a two-phase motor, and therefore, th

32、eir conclusions cannot applied directly to our situation, where a three-phase motor will be considered. Here, we give a more elegant analysis for stabilizing stepper motors, where no complex mathematical manipulation is needed. In this analysis, a dq model of stepper motors is used. Because two-phas

33、e motors and three-phase motors have the same qd model and therefore, the analysis is valid for both two-phase and three-phase motors. Up to date, it is only recognized that the modulation method is needed to suppress the midfrequency oscillation. In this paper, it is shown that this method is not o

34、nly valid to improve mid-frequency stability, but also effective to improve high-frequency stability.2. Dynamic Model of Stepper MotorsThe stepper motor considered in this paper consists of a salient stator with two-phase or threephase windings, and a permanent-magnet rotor. A simplified schematic o

35、f a three-phase motor with one pole-pair is shown in Figure 1. The stepper motor is usually fed by a voltage-source inverter, which is controlled by a sequence of pulses and produces square-wave voltages. Thismotor operates essentially on the same principle as that of synchronous motors. One of majo

36、r operating manner for stepper motors is that supplying voltage is kept constant and frequencyof pulses is changed at a very wide range. Under this operating condition, oscillation and instability problems usually arise.Figure 1. Schematic model of a three-phase stepper motor.A mathematical model fo

37、r a three-phase stepper motor is established using qd framereference transformation. The voltage equations for three-phase windings are given byva = Ria + L*dia /dt M*dib/dt M*dic/dt + dpma/dt ,vb = Rib + L*dib/dt M*dia/dt M*dic/dt + dpmb/dt ,vc = Ric + L*dic/dt M*dia/dt M*dib/dt + dpmc/dt ,where R

38、and L are the resistance and inductance of the phase windings, and M is the mutual inductance between the phase windings. _pma, _pmb and _pmc are the flux-linkages of thephases due to the permanent magnet, and can be assumed to be sinusoid functions of rotor position _ as followpma = 1 sin(N),pmb =

39、1 sin(N 2 QUOTE /3),pmc = 1 sin(N - 2 QUOTE /3),where N is number of rotor teeth. The nonlinearity emphasized in this paper is represented by the above equations, that is, the flux-linkages are nonlinear functions of the rotor position.By using the q; d transformation, the frame of reference is chan

40、ged from the fixed phase axes to the axes moving with the rotor (refer to Figure 2). Transformation matrix from the a; b; c frame to the q; d frame is given by 8For example, voltages in the q; d reference are given byIn the a; b; c reference, only two variables are independent (ia C ib C ic D 0); th

41、erefore, the above transformation from three variables to two variables is allowable. Applying the abovetransformation to the voltage equations (1), the transferred voltage equation in the q; d frame can be obtained asvq = Riq + L1*diq/dt + NL1id + N1,vd=Rid + L1*did/dt NL1iq, (5)Figure 2. a, b, c a

42、nd d, q reference frame.where L1 D L CM, and ! is the speed of the rotor.It can be shown that the motors torque has the following form 2T = 3/2N1iqThe equation of motion of the rotor is written asJ*d/dt = 3/2*N1iq Bf Tl ,where Bf is the coefficient of viscous friction, and Tl represents load torque,

43、 which is assumed to be a constant in this paper.In order to constitute the complete state equation of the motor, we need another state variable that represents the position of the rotor. For this purpose the so called load angle _ 8 is usually used, which satisfies the following equationD/dt = 0 ,w

44、here !0 is steady-state speed of the motor. Equations (5), (7), and (8) constitute the statespace model of the motor, for which the input variables are the voltages vq and vd. As mentioned before, stepper motors are fed by an inverter, whose output voltages are not sinusoidal but instead are square

45、waves. However, because the non-sinusoidal voltages do not change the oscillation feature and instability very much if compared to the sinusoidal case (as will be shown in Section 3, the oscillation is due to the nonlinearity of the motor), for the purposes of this paper we can assume the supply vol

46、tages are sinusoidal. Under this assumption, we can get vq and vd as followsvq = Vmcos(N) ,vd = Vmsin(N) ,where Vm is the maximum of the sine wave. With the above equation, we have changed the input voltages from a function of time to a function of state, and in this way we can represent the dynamic

47、s of the motor by a autonomous system, as shown below. This will simplify the mathematical analysis.From Equations (5), (7), and (8), the state-space model of the motor can be written in a matrix form as follows = F(X,u) = AX + Fn(X) + Bu , (10)where X D Tiq id ! _UT , u D T!1 TlUT is defined as the

48、 input, and !1 D N!0 is the supply frequency. The input matrix B is defined byThe matrix A is the linear part of F._/, and is given byFn.X/ represents the nonlinear part of F._/, and is given byThe input term u is independent of time, and therefore Equation (10) is autonomous.There are three paramet

49、ers in F.X;u/, they are the supply frequency !1, the supply voltage magnitude Vm and the load torque Tl . These parameters govern the behaviour of the stepper motor. In practice, stepper motors are usually driven in such a way that the supply frequency !1 is changed by the command pulse to control t

50、he motors speed, while the supply voltage is kept constant. Therefore, we shall investigate the effect of parameter !1.3. Bifurcation and Mid-Frequency OscillationBy setting ! D !0, the equilibria of Equation (10) are given asand is its phase angle defined by = arctan(1L1/R) . (16) Equations (12) an

51、d (13) indicate that multiple equilibria exist, which means that these equilibria can never be globally stable. One can see that there are two groups of equilibria as shown in Equations (12) and (13). The first group represented by Equation (12) corresponds to the real operating conditions of the mo

52、tor. The second group represented by Equation (13) is always unstable and does not relate to the real operating conditions. In the following, we will concentrate on the equilibria represented by Equation (12).附錄資料：不需要的可以自行刪除地下連續(xù)墻施工工藝標準1、范圍本工藝適用于工業(yè)與民用建筑地下連續(xù)墻基坑工程。地下連續(xù)墻是在地面上采用一種挖槽機械，沿著深開挖工程的周邊軸線，在泥漿護壁條

53、件下，開挖出一條狹長的深槽，清槽后，在槽內(nèi)吊放鋼筋籠，然后用導管法灌筑水下混凝土筑成一個單元槽段，如此逐段進行，在地下筑成一道連續(xù)的鋼筋混凝土墻壁，作為截水、防滲、承重、擋水結構。本法特點是：施工振動小，墻體剛度大，整體性好，施工速度快，可省土石方，可用于密集建筑群中建造深基坑支護及進行逆作法施工，可用于各種地質(zhì)條件下，包括砂性土層、粒徑50mm以下的砂礫層中施工等。適用于建造建筑物的地下室、地下商場、停車場、地下油庫、擋土墻、高層建筑的深基礎、逆作法施工圍護結構，工業(yè)建筑的深池、坑；豎井等。2、施工準備2.1材料要求2.1.1水泥用32.5號或42.5號普通硅酸鹽水泥或礦渣硅酸鹽水泥，要求新

54、鮮無結塊。2.1.2砂宜用粒度良好的中、粗砂，含泥量小于5%。2.1.3石子宜采用卵石，如使用碎石，應適當增加水泥用量及砂率，以保證坍落度及和易性的要求。其最大粒徑不應大于導管內(nèi)徑的16和鋼筋最小間距的14，且不大于40mm。含泥量小于2%。2.1.4外加劑可根據(jù)需要摻加減水劑、緩凝劑等外加劑，摻入量應通過試驗確定。2.1.5鋼筋按設計要求選用，應有出廠質(zhì)量證明書或試驗報告單，并應取試樣作機械性能試驗，合格后方可使用。2.1.6泥漿材料泥漿系由土料、水和摻合物組成。拌制泥漿使用膨潤土，細度應為200250目，膨潤率510倍，使用前應取樣進行泥漿配合比試驗。如采取粘土制漿時，應進行物理、化學分析

55、和礦物鑒定，其粘粒含量應大于50%，塑性指數(shù)大于20，含砂量小于5%，二氧化硅與三氧化鋁含量的比值宜為34。摻合物有分散劑、增粘劑(CMC)等。外加劑的選擇和配方需經(jīng)試驗確定，制備泥漿用水應不含雜質(zhì)，pH值為79。2.2主要機具設備2.2.1成槽設備有多頭鉆成槽機、抓斗式成槽機、沖擊鉆、砂泵或空氣吸泥機(包括空壓機)、軌道轉盤等2.2.2混凝土澆灌機具有混凝土攪拌機、澆灌架(包括儲料斗、吊車或卷揚機)、金屬導管和運輸設備等。2.2.3制漿機具有泥漿攪拌機、泥漿泵、空壓機、水泵、軟軸攪拌器、旋流器、振動篩、泥漿比重秤、漏斗粘度計、秒表、量筒或量杯、失水量儀、靜切力計、含砂量測定器、pH試紙等。2

56、.2.4槽段接頭設備有金屬接頭管、履帶或輪胎式起重機、頂升架(包括支承架、大行程千斤頂和油泵等)或振動拔管機等。2.2.5其他機具設備有鋼筋對焊機，彎曲機，切斷機，交、直流電焊機，大、小平鍬，各種扳手等。2.3作業(yè)條件、2.3.1在工程范圍內(nèi)鉆探，查明地質(zhì)、地層、土質(zhì)以及水文情況，為選擇挖槽機具、泥漿循環(huán)工藝、槽段長度等提供可靠的技術數(shù)據(jù).。同時進行鉆探，摸清地下連續(xù)墻部位的地下障礙物情況。2.3.2按設計地面標高進行場地平整，拆遷施工區(qū)域內(nèi)的房屋、通訊、電力設施以及上下水管道等障礙物，挖除工程部位地面以下m內(nèi)的地下障礙物。施工場地周圍設置排水系統(tǒng)。2.3.3根據(jù)工程結構、地質(zhì)情況及施工條件制

57、定施工方案，選定并準備機具設備，進行施工部署、平面規(guī)劃、勞動配備及劃分槽段；確定泥漿配合比、配制及處理方法，編制材料、施工機具需用量計劃及技術培訓計劃，提出保證質(zhì)量、安全及節(jié)約等的技術措施。2.3.4按平面及工藝要求設置臨時設施，修筑道路，在施工區(qū)域設置導墻；安裝挖槽、泥漿制配、處理、鋼筋加工機具設備；安裝水電線路；進行試通水、通電、試運轉、試挖槽、混凝土試澆灌。3、操作工藝3.1工藝流程(圖3.1)圖3.1多頭鉆施工及泥漿循環(huán)工藝3.2導墻設置3.2.1在槽段開挖前，沿連續(xù)墻縱向軸線位置構筑導墻，采用現(xiàn)澆混凝土或鋼筋混凝土澆3.2.2導墻深度一般為12m，其頂面略高于地面50100mm，以防

58、止地表水流入導溝。導墻的厚度一般為100200mm，內(nèi)墻面應垂直，內(nèi)壁凈距應為連續(xù)墻設計厚度加施工余量(一般為4060mm)。墻面與縱軸線距離的允許偏差為10mm，內(nèi)外導墻間距允許偏蓋5mm，導墻頂面應保持水平。3.2.3導墻宜筑于密實的粘性土地基上。墻背宜以土壁代模，以防止槽外地表水滲入槽內(nèi)。如果墻背側需回填土時，應用粘性土分層夯實，以免漏漿。每個槽段內(nèi)的導墻應設一溢漿孔。3.2.4導墻頂面應高出地下水位1m以上，以保證槽內(nèi)泥漿液面高于地下水位0.5m以上，且不低于導墻頂面0.3m。3.2.5導墻混凝土強度應達到70%以上方可拆模。拆模后，應立即將導墻間加木支撐至槽段開挖拆除。嚴禁重型機械通

59、過、停置或作業(yè)，以防導墻開裂或變形。3.3泥漿制備和使用3.3.1泥漿的性能和技術指標，應根據(jù)成槽方法和地質(zhì)情況而定，一般可按表3.3.1采用。泥漿性能指標表3.3.1項目性能指標檢查方法一般地層軟弱土層密度粘度膠體率穩(wěn)定性失水量pH值泥皮厚度靜切力(1min)含砂量1.041.25kgL1822s95%0.05gcm330mL30min101.53.0mm30min1020mgcm298%0.02gcm320mL30min891.01.5mm30min2050mgcm24%泥漿密度秤500700mL漏斗法100mL量杯法500mL量筒或穩(wěn)定計失水量儀pH試紙失水量儀靜切力計含砂量測定器注：1

60、.密度：表中上限為新制泥漿，下限為循環(huán)泥漿。一般采用膨潤土泥漿時，新漿密度控制在1.041.05；循環(huán)程中的泥漿控制在1.251.30；對于松散易坍地層，密度可適當加大。澆灌混凝土前槽內(nèi)泥漿控制在1.151.25，視土質(zhì)情況而定；2.成槽時，泥漿主要起護壁作用，在一般情況下可只考慮密度、粘度、膠體率三項指標；3.當存在易塌方土層(如砂層或地下水位下的粉砂層等)或采用產(chǎn)生沖擊、沖刷的掘削機械時，應適當考慮，泥漿粘度，宜用2530s。3.3.2在施工過程中應加強檢查和控制泥漿的性能，定時對泥漿性能進行測試，隨時調(diào)泥漿配合比，做好泥漿質(zhì)量檢測記錄。一般作法是：在新漿拌制后靜止24h，測一次全項(含砂