LTE關(guān)鍵技術(shù)之OFDM分析-外文翻譯

1、201x屆本科畢業(yè)設(shè)計(jì)（外文翻譯） 學(xué) 院: 專(zhuān) 業(yè): 姓 名: 學(xué) 號(hào): 指導(dǎo)教師： 完成時(shí)間： 二0四年三月 LTE的多址接入技術(shù) LTE的多址接入 OFDM傳輸 正交頻分復(fù)用（OFDM ）是一種多載波傳輸技術(shù)，已被采納為 3gpplong長(zhǎng) 期演化（LTE）的下行鏈路傳輸方案，也可用于其他幾個(gè)無(wú)線技術(shù)，例如：wimax 和DVB廣播技術(shù)。它的特點(diǎn)是在一個(gè)頻域內(nèi)分布著許多帶有間隔的子載波 f=1/Tu其中，Tu是每個(gè)子載波的調(diào)制符號(hào)時(shí)間。如圖 2-1所示，OFDM子載 波間隔”。 OFDM的傳輸是基于塊的。每個(gè) OFDM符號(hào)間隔之間，調(diào)制符號(hào)是并行發(fā) 送的。調(diào)制符號(hào)可以通過(guò)調(diào)制字母表得到，

2、如 QPSK, 16QAM或64QAM，對(duì)于 3GPP組織LTE，子載波間隔是相等的為15 kHz。另一方面，子載波的數(shù)目取決 于傳輸帶寬，在一個(gè)10MHZ的頻譜分配下，600個(gè)子載波可以有序傳輸。當(dāng)然， 帶寬減小了，子載波數(shù)目也相應(yīng)減少，帶寬增加了，子載波數(shù)目也相應(yīng)增加。 Af = 1/Tu * 圖2-1 OFDM子載波間隔 在OFDM傳輸時(shí)，物理資源經(jīng)常被描述成一個(gè)時(shí)域 一頻域的網(wǎng)格坐標(biāo)圖。 在這個(gè)坐標(biāo)圖里一列對(duì)應(yīng)一個(gè) OFDM子載波，一行對(duì)應(yīng)一個(gè) OFDM子載波。如 圖2-2所示，OFDM時(shí)頻網(wǎng)格”。 盡管子載波的頻譜有重疊，但在理想情況下，是對(duì) OFDM子載波解調(diào)后不 引起任何干擾的，

3、這是因?yàn)閷?duì)每一個(gè)子載波間隔的特殊選擇， 讓它等于相應(yīng)的解 調(diào)符號(hào)率。 time 圖2-2 OFDM時(shí)頻網(wǎng)格 subcarrier spacing = Af Frequency = 3f Frequency = 1 OFDM symbol Frequency = f 以一定的頻率fs= N x 進(jìn)行采樣的OFDM信號(hào)，是該size-N的逆離散傅立葉 變換（IDFT）的調(diào)制符號(hào)塊ao, ai,.aN-i。因此，OFDM調(diào)制可以通過(guò)IDFT處理再 到數(shù)字-模擬的轉(zhuǎn)換來(lái)實(shí)現(xiàn)。（見(jiàn)圖2-3，“OFD碉制”）。在實(shí)際中，OFDM調(diào)制 是以快速傅立葉反變換（IFFT）方式實(shí)現(xiàn)簡(jiǎn)單和快速的處理，通過(guò)選擇IDF

4、T size N等于2m （m為整數(shù)）。在接收端，對(duì)接收信號(hào)以fs= N X的頻率采樣，高效的 FFT處理是用來(lái)實(shí)現(xiàn)OFDM的解調(diào)和檢索調(diào)制符號(hào)塊ao, ai,.aN-i。（參見(jiàn)圖2-4, OFDM解調(diào)” Frequency lo M且未使用的輸入(N-M )設(shè)置為零。和OFDM 一樣，每個(gè)傳輸塊插入一個(gè)循環(huán)前綴。 bit stream No 卡n-telI in悴恂- different men ue different njbcarnen SizeN Localized Rejected in 36PP oim poor channel est. performance 汕d seniit

5、lvitv to frequency offset From DFT (Size From DFT (Size M SizeN I F 圖2-8 DFT的OFDM信號(hào)的產(chǎn)生 與圖2-8,“ DFT勺OFDM信號(hào)生成”相比，基于IFFT OFDM調(diào)制的實(shí)現(xiàn)，很 顯然，DFTS-OFDM可以看作是OFDM調(diào)制之前的DFT運(yùn)算。如果DFT的M的大 小等于IDFT的N的大小，那么級(jí)聯(lián)DFT和IDFT的塊圖2-8 DFT的OFDM信號(hào)生成” 將完全抵消。如果M小于N且IDFT的剩余輸入被設(shè)置為零，則IDFT的輸出將是一 個(gè)低功率變化的信號(hào)，類(lèi)似于一個(gè)單載波信號(hào)。此外，不同塊大小為m的瞬時(shí)帶 寬發(fā)送的信號(hào)

6、可以是多種多樣的，允許靈活的帶寬分配。 與DFTS-OFDM的主要好處想比，多載波傳輸方案，如 OFDM,減少變化的瞬 時(shí)發(fā)射功率，對(duì)提高功率放大器效率是可能的。功率的變化一般根據(jù)測(cè)得的峰值 平均功率比（PRPA）來(lái)判斷。定義為在峰值功率一個(gè) OFDM符號(hào)的平均信號(hào)功 率的歸一化。對(duì)于DFTS-OFDM,PRPA明顯降低，相比OFDM,再考慮到移動(dòng)終端 的電源能力，這種傳輸技術(shù)在上行鏈路的傳輸中是非常有用的。 DFTS-OFDM信號(hào)解調(diào)的基本原理如圖2-9所示，DFT的OFDM解調(diào)”。這些 操作和圖2-9 DFT的OFDM解調(diào)”基本上是相反的。即size-n離散傅里葉變換處理 中，和接受信號(hào)不

7、對(duì)應(yīng)的頻率采樣會(huì)被移除。 Time Io frequiency domain confers Ion 圖 2-9 DFTS OFDM 調(diào)希 9 LTE multiple access tech niq ues LTE multiple access OFDM tran smissio n Orthogonal Frequency Division Multiplexing (OFDM) is a multicarrier transmission tech nique that has bee n adopted as the dow nli nk tran smissi on scheme

8、for the 3GPP Lon g-Term Evolutio n (LTE) and is also used for several other radio tech no logies, e.g. WiMAX and the DVB broadcast tech no logies. It is characterized by a tight frequency-domain packing of the subcarriers with a subcarrier spac ingf = 1/Tu, where Tu is the per-subcarrier modulatio n

9、-symbol time. (SeeFigure 2-1,“ OFDM subcarrier spacing ” OFDM tran smissi on is block-based. During each OFDM symbol in terval, modulati on symbols are transmitted in parallel. The modulation symbols can be from any modulation alphabet, such as QPSK, 16QAM, or 64QAM. For 3GPP LTE, the basic subcarri

10、er spaci ng equals 15 kHz. On the other hand, the number of subcarriers depends on the transmission bandwidth, with in the order of 600 subcarriers in case of operation in a 10 MHz spectrum allocation and corresp on di nglyfewr/more subcarriers in case of smaller/larger overall tran smissi on bandwi

11、dths.rI I I I I i I I I Figure 2-1 OFDM subcarrier spaci ng sub匚忌ier spacing = f ission is often illustrated as a s to one OFDM symbol (time) and a ncy = 3 f 丁 , pite the .fact that lhe spectrum of n eighbor subcarriers do overlap, theOFDM subcarrie+sdo not cause any in terfere nee to each other aft

12、er demodulati on due tothe specific choice of a subcarrier spacing f equal to the modulation symbol - rate. The physica time-frequer row corres time-frequer In the ideal case, reso in cy gri vh ponds tojn case of OFDM iere . a column cai ie OFDMsubca i血stated . in .(seFigure 2-2,“卩罔冊(cè) I 11I Hi W ion

13、Figure 2-2 OFDM time-freque ncy grid Af= 1/Tu * * Frequen匚y - Af * I Time to frequonKy dGcnafn conver-&ioii MO a)FDM Signal samplqd aja rat FratsfoBmkDFTpof the bockfn Nf is laYionsymbBls modulation can be implemented by digital-to-analog conversion (seeFigure 2-3, means he siz OFD 4 e-N nvese Discr

14、ete Fourier ao, ?1,下壯1 Thus OFD * ocessing followed by MmOdulation| ”n) dractic4,the OFDM modulati on can be impleme nted by mea ns of In verse Fast Fourier Tran sform (IFFT) easy and fast process ing, by selecti ng the IDFT sizN equal to 2m for some in tegerm. At the receiver, by sampli ng the rece

15、ived sig nal at the rates = N xf, efficient FFT processing is used to achieve OFDM demodulation and retrieve the block of modulation symbols ao, ai,.aN-i( seeFigure 2-4,“OFDM demodulation. ” ) Figure 2-3 OFDM modulation 皿呻呻怖伽Figure 2-4 OFDM demodulation As men ti oned above, an un corrupted OFDM sig

16、 nal can be demodulated without any in terfere nee betwee n subcarriers. However, i n case of a time-dispersive cha nn el (such asmultipath radio channels), the orthogonality between the subcarriers is lost, causing In ter Symbol In terfere nce (ISI). The reas on for this is that the demodulator cor

17、relati on interval for one path will overlap with the symbol boundary of a different path (see Figure 2-5, “ Time dispersion and corresponding received signal” Figure 2-5 Time dispersi on and corresp onding received sig nal Cyclic prefix permits to facilitate demodulation k i 1V M I To deal wit robl

18、em and make an OFDM sig nal truly insen sitive to 種門(mén) el, so-called Cyclic Prefix inserti on is typically used in transmiss in. As illustrated in( seeFigure 2-6；yclic Prefix ：,”)clic-prefiX/ nsertio n implies that the last part of the OFDM symbol (the last Ncp symbols) is copieda nd in serted at the

19、begi nning of the OFDM block, increasi ng thus the len gth of theOFDM symbol from Tu to Tu + Tcp, where Tcp = NcplTJ is the len gth of the cyclic prefix.心忖/( ij The OFDM symbol rate as is reduced as a con sequence. Thus, subcarrier orthogonality is preserved in case of a ti-dispersivechan the time d

20、ispersionis shorter than the cyclic-prefix length. Figure 2-6 Cyclic Prefix in sertio n i1 Jil fi li EXiMfci long as the spa n o.一 4 Mrililiiiil mb eNode-B Radi M and where the(N-M) unu sed in puts of the IDFT are set to zero. Also similar to OFDM, a cyclic prefix is inserted for each transmitted bl

21、ock. Figure 2-8 DFTS OFDM sig nal ge neratio n Tim*? in fmifimncy cluingn Comparing (seeFigure 2-8, “ DFTS OFDM signgael neration ”w)i,th the IFFT-based implementation of OFDM modulation, it is obvious that DFTS-OFDM can alternatively be seen as OFDM modulation preceded by a DFT operation. If the DF

22、T size M equals the IDFT size N, the cascaded DFT and IDFT blocks of (see Figure 2-8, “DFTS OFDM signal generationwill ”co)m, pletely cancel out each other. However, if M is smaller than N and the remaining inputs to the IDFT are set to zero, the output of the IDFT will be a signal with low power va

23、riations, similar to a single-carrier signal. Besides, by varying the block size M the instantaneous bandwidth of the transmitted signal can be varied, allowing for flexible-bandwidth assignment.The main benefit of DFTS-OFDM, compared to a multi-carrier transmission scheme suchas OFDM, is reduced variations in the instantaneous transmit power, implying the possibility for increased power-amplifier efficiency. The power variations a